Rationale for a Selectable Performance Filter and Critical Characteristics Thereof
Surface Acoustic Wave (SAW) programmable transversal filters with a limited degree of selectable performance are well known in the art. A fixed input interdigitated transducer (hereinafter IDT) and an output array if individual finger taps are disposed on each SAW substrate of such filters. Selectable performance is attained by connecting each of the output finger taps through a corresponding array of variable gain amplifiers, and forming the overall filter output as the summed output of the amplifier array. Several significant limitations, disadvantages and shortcomings have been encountered with such prior art filters. Center frequency tunability is restricted to essentially the passband of the fixed input IDT. Bandwidth tunability is accomplished in a cumbersome manner by varying the effective length of the output array, and is limited to those bandwidths associated with the output array delay length. Furthermore, the weighting circuit must perform a complex multiplication to provide unaliased center frequency tuning.
Communications receivers are often faced with the problem of trying to receive a weak communications signal in the presence of one or more strong interfering signals. Whereas the receiver may have sufficient sensitivity to receive the weak signal against a “quiet” background, when the background includes one or more strong interfering signals, the receiver will be desensitized to a level determined by the strength of the largest interfering signal and the dynamic range of the receiver. In these situations, it is highly desirable to include some sort of RF filtering at the receiver front-end in order to attenuate the interfering signals prior to the first gain stage. The critical RF parameters of the selectable performance front-end filter can be derived through a simple analysis of a generic receiver circuit. The analysis begins by considering the signal set as illustrated in FIG. 1. Desired signal 1 and undesired higher power interference signal 2 are present simultaneously but at different frequencies. These signals are applied to a generic, wide-open receiver as shown in FIG. 2. The major components of the receiver are antenna 10, RF amplifier 11, detector 12, IF amplifier 13 and comparator 14. The input to RF amplifier 11 is considered to be hard limited to a maximum power level Psat. RF amplifier 11 is characterized by gain G, noise figure F and noise power bandwidth B0. Detector 12 is characterized by a tangential signal sensitivity TSS.
In a “quiet” RF enviromment, one can use an amount of gain G given byG=[TSS−(N+F)]  (1)for maximum system sensitivity to weak signals. With a strong interference signal of P>(Psat−G), conventional automatic gain control (AGC) circuitry will reduce the system sensitivity by 1 dB for each 1 dB of P>(Psat−G) up to the hard limit, i.e., there is a potential loss in sensitivity up to a value of G. This process is illustrated in FIG. 3. FIG. 3(a) presents the power levels of desired signal 1 and undesired higher power interference signal 2 at the receiver input. These signals are compared to the thermal noise input to the receiver given by N=kTB0, the thermal noise at the input to the detector given by N+F, the largest input signal level that maintains maximum signal sensitivity given by Psat−G(ideal), the TSS of the detector and the hard limit Psat. Given that the power level of undesired higher power interference signal 2 exceeds Psat−G(ideal), the AGC will reduce the gain to a value such that Psat−G(actual) is equal to the input power level of interference signal 2. FIG. 3(b) presents the resulting power levels of desired signal 1 and undesired interference signal 2 at the detector. These signals are compared to the same levels as in FIG. 3(a), along with the minimum detectable signal level given by TSS-G(actual). Both desired signal 1 and undesired signal 2 are increased by the value of G(actual), however this reduced value of gain is not enough to raise desired signal 1 being above the minimum detectable signal level.
One can add an RF filter 25 to the receiver front-end as illustrated in FIG. 4. The other major components of the receiver are as before antenna 20, RF amplifier 21, detector 22, IF amplifier 23 and comparator 24. RF filter 25 is characterized by bandwidth B, insertion loss I, and sidelobe (rejection) level S. The power of the undesired interference signal 2 is reduced by S dB. If the sidelobe (rejection) level is greater than the gain, that is, for S>G, the undesired interference signal 2 does not desensitize the receiver. At the same time, the insertion loss of RF filter 25 reduces the power in desired signal 1 by I dB, which for a constant noise floor, reduces the receiver sensitivity by I dB. However, the noise floor is increased by a factor of (F/I) arising from noise in RF amplifier 21 and decreased by a factor of 10log(B/B0) since the input thermal noise is band-limited by RF filter 25. If the decrease in noise floor due to the bandwidth reduction exceeds the noise increase due to the filter noise figure (≈1/I), then additional RF gain can be added for increased sensitivity. These considerations are illustrated in FIG. 5, wherein the signal power levels at the detector are illustrated for the receiver with front-end filter of FIG. 4. Desired signal 1 is first reduced by insertion loss I from an initial level of Psoi and then increased by gain G to a net power level of Psoi+G−I at the detector. In contrast, undesired interference signal 2 is first reduced by sidelobe (rejection) level S from an initial level of Psoi and then increased by gain G to a net power level of Psnoi+G−S at the detector. Thus, compared to a wide open system with RF bandwidth B0 as illustrated in FIG. 4, the sensitivity in a quiet RF environment is degraded by I+(F/I)−10log(B/B0), but the sensitivity in the presence of strong interference is improved by as much as G−[I+(F/I)−10log(B/B0)] provided that S>G. The net result is a figure of merit for the system performance improvement resulting from RF filter 25 essentially given by sidelobe level S, which establishes how well the desensitization is obviated, minus the insertion loss I, which establishes how much the quiescent performance is degraded.
History and Background of Programmable SAW Transversal Filters
The above considerations, of course, presuppose that the filter center frequency and bandwidth are chosen so as to include desired signal 1 in the passband, and to exclude the undesired interference signal 2. Taking all of these factors into account, if RF filter 25 included a selectable center frequency, a selectable bandwidth, a minimum insertion loss, a maximum sidelobe (rejection) level and a selectable filter transfer function (phase/magnitude profile) the filter would seem to overcome the disadvantages, shortcomings and limitations of prior art devices. The ability of RF filter 25 to meet the performance goals depends in part on the choice of filter topology, and in part upon the technology used to implement RF filter 25. Prior developments in the field of programmable transversal filters based on surface acoustic wave devices and closely related fields that provide useful guidance on these choices are reviewed in “Design of a Selectable Performance Front End Filter Using Acoustic Surface Wave Resonators,” by R. Pastore, J. A. Kosinski, W. N. Porter, and H. L. Cui, in Proceedings of the 1997 IEEE International Frequency Control Symposium, May 1997, pp. 858–866.
Filter and device topologies considered previously include IDT arrays, filter banks, variable wave velocity devices, dispersive delay devices, convolvers, correlators, matched filters and transversal filters. These devices have been implemented primarily using SAW technology. A device based upon acoustic charge transport (ACT) technology is disclosed in U.S. Pat. No. 5,225,798 titled “Programmable Transversal Filter” issued Jul. 6, 1993 to B. J. Hunsinger and J. E. Bales. Devices based upon charge coupled device (CCD) technology are disclosed in U.S. Pat. No. 4,612,522 titled “Mask Programmable Charge Coupled Device Transversal Filter” issued Sep. 16, 1986 to R. H. Dyck, and U.S. Pat. No. 4,034,199 titled “Programmable Analog Transversal Filter” issued Jul. 5, 1977 to D. R. Lampe, M. H. White, and J. H. Mims.
IDT arrays have been used to obtain selectable bandwidth. However, this technique requires unrealistically large delay paths to obtain narrow filter bandwidths. The filter bank approaches have achieved good electrical performance over limited ranges of center frequency and bandwidth. However, broadband tunability using this technique would require a substantial number of channels, also leading to prohibitively large devices.
Variable wave velocity has been used to obtain selectable delay time. However, this tuning is limited to a small fraction of the nominal wave velocity. Selectable center frequency also has been demonstrated using structures based on dispersive devices. However these structures are somewhat complicated and have inherent blanking time and bandwidth limitations.
Transversal filters have demonstrated selectable bandwidth and selectable center frequency as disclosed in U.S. Pat. No. 5,387,887 titled “Miniature Digitally Controlled Programmable Transversal Filter Using LSI GaAs Integrated Circuits” issued Feb. 7, 1995 to D. E. Zimmerman, J. W. Colver, and C. M. Panasik. To date, however, this selectability has been over a limited range determined in part by the use of a conventional input IDT structure. Promising results have been reported in regards to insertion loss (10 dB) and dynamic range (>70 dB), with less promising results reported for sidelobe levels (35 dB).
The full range of physical implementations have been reported, including discrete piezoelectric devices, hybrid devices using thin film piezoelectrics deposited on semiconducting substrates, and monolithic implementations using piezoelectric semiconductors.
Silicon, silicon-on-sapphire (SOS), and gallium arsenide have been used to implement the required tap weight and control circuits. Silicon has advantages with respect to cost and processing, while gallium arsenide has advantages with respect to critical circuit parameters.
Both passive and active tap weight control elements have been implemented. Active tap weight control elements have been based primarily on FET's, with a lone reference to bipolar technology. An alternative approach for tap weight control using fixed gain amplifiers and switch arrays is disclosed in Kosinski U.S. Pat. No. 6,492,884, entitled “Programmable Transversal Filter,” issued on Dec. 10, 2002.
A wide variety of piezoelectric materials have been used to date, including piezoelectric ceramics, piezoelectric semiconductors, piezoelectric thin films (polycrystalline) and conventional single crystals. The most popular material to date has been single crystal lithium niobate.
Operating Principles of the Prior Art SAW Filters
The performance goals for RF filter 25 can be obtained readily with an ideal transversal filter 30 as illustrated in FIG. 6. In transversal filter 30, the input signal is applied to an input terminal 31 and propagates through a series of N delay elements 32. The time delay through each delay element 32 is denoted τn. The signal is sampled between the delay elements 32, with each such sample following a path through one of N corresponding gain or weight control element 33 and thence to a summation element 34. The filter output 35 thus is formed as a weighted sum of the N delayed signal samples. Transversal filter 30 is, in theory, capable of producing an arbitrary transfer function
                              H          ⁡                      (            f            )                          =                              ∑                          n              =              1                        N                    ⁢                                    a              n                        ⁢                                          ⅇ                                                      -                    j2                                    ⁢                                                                          ⁢                  π                  ⁢                                                                          ⁢                  f                  ⁢                                                                          ⁢                                      τ                    n                                                              .                                                          (        2        )            
The implementation of transversal filter 30 as a tapped delay line using conventional SAW technology is illustrated in FIG. 7. SAW transversal filter 40 is comprised of input IDT 42 and output IDT 43 deposited on the surface of piezoelectric substrate 41. Each IDT is composed of interdigitated fingers 45 alternately connected to busbars 44. The delay times τn are determined by the locations of the IDT fingers 45 along the propagation direction 46 and the SAW propagation velocity. The individual tap weights an are established by techniques such as overlap, withdrawal, or phase-reversal weighting, and are implemented as fixed weights determined by the geometry of the IDT fingers 45. The summation operation is performed by the busbars 44 connecting the IDT fingers 45.
The SAW device geometry is primarily defined by the finger locations xn of the fingers 45 of input IDT 42 and by the finger locations ym of the fingers 45 of output IDT 43 as defined in FIG. 7. The transfer function for SAW transversal filter 40 for the case of unapodized (equal finger length) IDTs can be written
                                          H            ⁡                          (              f              )                                =                                    ∑                              n                =                1                            N                        ⁢                                          I                n                            ⁢                              ⅇ                                                      -                    j                                    ⁢                                                            2                      ⁢                      π                      ⁢                                                                                          ⁢                      f                                                              v                      s                                                        ⁢                                      x                    n                                                              ⁢                                                ∑                                      m                    =                    1                                    M                                ⁢                                                      J                    m                                    ⁢                                      ⅇ                                          j                      ⁢                                                                        2                          ⁢                          π                          ⁢                                                                                                          ⁢                          f                                                                          v                          s                                                                    ⁢                                              y                        m                                                                                                                                ,                            (        3        )            where In and Jm represent the input and output IDT tap weights respectively. For unapodized IDTs, In=Jm=1 and H(f) is of the formH(f)=Hin(f) Hout(f)  (4)
Note that generic transversal filter 30 provides for a baseband or lowpass filter response, whereas SAW transversal filter 40 produces a bandpass response. This is a result of the polar nature of the piezoelectric effect, which requires that IDT fingers 45 have alternating polarity, implemented via the alternating connections of fingers 45 to busbars 44. Consequently, the SAW tap weights incorporate a factor of
                                                        (                              -                1                            )                        n                    =                                    cos              ⁡                              (                                                      2                    ⁢                    π                    ⁢                                                                                  ⁢                                          f                      0                                        ⁢                                          x                      n                                                                            v                    s                                                  )                                      =                          cos              ⁡                              (                                  2                  ⁢                  π                  ⁢                                                                          ⁢                                      f                    0                                    ⁢                                                                          ⁢                                      τ                    n                                                  )                                                    ,                            (        5        )            which is equivalent to frequency translation of the baseband response to a carrier frequency f0. This implicit frequency translation mechanism can be extended to essentially arbitrary frequency translation to fc by incorporation of an additional factor of exp[j2π(fc−f0)] in the IDT tap weights. This principle is the basis of a tunable output IDT as employed in the partially programmable SAW transversal filter disclosed by D. E. Zimmerman, J. W. Colver, and C. M. Panasik in U.S. Pat. No. 5,387,887. In the tunable IDT, the tap weights are not established as fixed values by the by the geometry of the IDT metallization using conventional overlap, withdrawal, or phase-reversal weighting techniques and connection to a common set of busbars. Rather, the tunable IDT fingers are formed as independent conductive stripes, and the tap weights are established by circuit elements interposed between the tunable IDT fingers and the summation circuit in order to effect the required tuning factor of exp[j2π(fc−f0)] in the IDT tap weights.
The programmable SAW transversal filters reported to date such as that of U.S. Pat. No. 5,387,887 are only partially programmable, in that they use conventional, fixed input IDTs in combination with tunable output IDTs. This partially programmable SAW transversal filter topology is illustrated schematically in FIG. 8. FIG. 8(a) illustrates the topology for the case of real tap weights, and FIG. 8(b) illustrates the topology for the case of complex tap weights. In FIG. 8(a), input signal 50 is applied to conventional IDT 51, which then generates an acoustic signal that passes to finger array 52. The electrical signals generated in finger array 52 are processed through tap weight network 53, and the output signal 55 is formed as the weighted sum via summation circuit 54. As shown in FIG. 8(b), complex tap weights are implemented using two parallel channels coupled through 90° hybrid couplers 56 at both input and output ports. In the SAW device implementation, input IDT 51 and finger array 52 are implemented as a piezoelectric device 60 as illustrated in FIG. 9. Piezoelectric device 60 is comprised of piezoelectric substrate 61, upon which are deposited conventional input IDT 62 and tunable output IDT 63. Conventional input IDT 62 is comprised of interdigitated fingers 65 alternately connected to busbars 64. Tunable output IDT 63 is comprised of independent fingers 66 each having a separate bonding pad 67 used to connect to the external tap weight circuitry. As in a conventional fixed SAW device, the partially programmable SAW transversal filter geometry is primarily defined by the finger locations xn of the fingers 65 of input IDT 62 and by the finger locations ym of the fingers 66 of output IDT 63 as defined in FIG. 9.
It is significant to note that, for the partially programmable SAW transversal filters considered to date, the periodicity of the fingers in both input and output IDTs is the same. That is, the input and output IDTs in the prior art share a common synchronous frequency. While promising in some aspects, the partially programmable SAW transversal filter topology of the prior art is inherently limited in meeting necessary performance goals such as selectable center frequency, selectable bandwidth, minimum insertion loss, maximum sidelobe (rejection) level and selectable filter transfer function (phase/magnitude profile). Thus there has been a long-felt need for a filter that meets these requirements and does not suffer from the disadvantages, shortcomings and limitations of the prior art devices.
Limitations of the Prior Art and Approaches Considered to Overcome the Limitations
A first limitation on center frequency tunability arises from the use of a fixed input IDT, in that low insertion loss can only be obtained for center frequencies falling within the input IDT coupling bandwidth. This limitation is illustrated in FIG. 10(a) for the case of complex tap weights implemented as illustrated in FIG. 8(b). The transfer function 70 of the fixed input IDT and the untuned transfer function 71 of the tunable output IDT share a common synchronous frequency 72. The transfer function 70 of the fixed input IDT is maximum at the synchronous frequency 72, and since the total transfer function of the filter is the product of the input IDT transfer function 70 and the output IDT transfer function 71, the untuned filter transfer function is essentially the same as the transfer function 71 of the untuned output IDT. However, the application of frequency tuning to the output IDT results in a new, tuned output IDT transfer function 74 centered at a tuned frequency 73 different from the synchronous frequency 72. In this case, the product of the input IDT transfer function 70 and the tuned output IDT transfer function 74 results in a tuned filter transfer function 75 that is an attenuated and distorted version of the transfer function 71 of the untuned output IDT. A tunable input IDT would overcome this limitation, thus creating a fully programmable SAW transversal filter. The advantages of such an arrangement as well as the case for complex tap weights are illustrated in FIG. 10(b). The untuned transfer function 76 of the tunable input IDT and the untuned transfer function 71 of the tunable output IDT share a common synchronous frequency 72. In the fully programmable SAW transversal filter, frequency tuning is applied to both input and output IDTs resulting in tuned input IDT transfer function 77 and tuned output IDT transfer function 74 both centered at a tuned frequency 73 different from the synchronous frequency 72. In this case, the product of the tuned input IDT transfer function 77 and the tuned output IDT transfer function 74 results in a tuned filter transfer function essentially the same as the tuned output transfer function 74 without the attenuation and distortion of the prior art partially programmable SAW transversal filter. However, it should be noted that the advantages illustrated in FIG. 10 accrue only for the case of complex tap weights, thus requiring a complicated dual channel hardware implementation.
The limitation inherent in single channel implementations of real tap weights is illustrated in FIG. 11 for both (a) partially programmable and (b) fully programmable SAW transversal filters. Whereas complex tap weights provide unambiguous frequency translation, the use of real tap weights results in aliasing of the tuned IDT passband. In the untuned partially programmable SAW transversal filter, the transfer function 80 of the fixed input IDT and the untuned transfer function 81 of the tunable output IDT share a common synchronous frequency 82. The untuned output IDT transfer function 81, and hence the untuned filter transfer function formed as the product of the input and output IDT transfer functions both feature a single passband. However, when frequency tuning is applied using real tap weights, the output IDT transfer function develops two allowed passbands 83 and 84, each of which is a copy of the original untuned output IDT transfer function 81 displaced symmetrically above and below the synchronous frequency to new passband center frequencies 85 and 86. In consequence, the tuned filter transfer function exhibits two allowed passbands 87 and 88, each an attenuated and distorted copy of the original untuned transfer function.
The situation is not significantly different for the simple fully programmable SAW transversal filter of FIG. 12. The simple fully programmable SAW transversal filter 100 is comprised of piezoelectric substrate 101, upon which are deposited tunable input IDT 102 and tunable output IDT 103. Tunable input IDT 102 is composed of independent fingers 105 each having a separate bonding pad 104 used to connect to the external input tap weight circuitry. Similarly, tunable output IDT 103 is also composed of independent fingers 106 each having a separate bonding pad 107 used to connect to the external output tap weight circuitry. As in conventional SAW devices and partially programmable SAW transversal filters, the fully programmable SAW transversal filter geometry is primarily defined by the finger locations xn of the fingers 105 of input IDT 102 and by the finger locations ym of the fingers 106 of output IDT 103 as defined in FIG. 12. It is significant to note that for the simple fully programmable SAW transversal filter as illustrated, the periodicity of the fingers in both input and output IDTs is the same. That is, the input and output IDTs in the prior art share a common synchronous frequency. As illustrated in FIG. 11(b), the substitution of a tunable input IDT with untuned transfer function 89 simply leads to aliased tuned input IDT passbands 90 and 91. The shift from partial to full programmability reduces the attenuation and distortion of the overall filter transfer function, but does not eliminate the limitation of passband aliasing in the simpler and hence inherently more desirable single channel device using real tap weights.
The prior art also suffers from limitations on selectability of the passband bandwidth. The bandwidth of the IDT passband is approximately 1/T where T is the delay time required for the acoustic wave to propagate across the IDT. A longer IDT has a longer delay time and hence a narrower bandwidth, while a shorter IDT has the opposite characteristics. By turning taps on and off, the active length of the tunable output IDT can be varied from as short as ½ wavelength to as long as (N−1)/2 wavelengths where N is the number of active taps. Thus, the maximum active length of the tunable output IDT limits the minimum bandwidth of the prior art filter.